Transmit method and system for Ka band transmissions

ABSTRACT

An improved method and system for generating quadrature phase shift keying signals for use in data transmission is provided. A pair of oscillators are slaved to the transmit frequency and produce two quadrature signal components with the same frequency but 90 degrees out of phase. The two signal components are carried to separate bi-phase switches by mirrored waveguides. Each bi-phase switch has a reflective waveguide coupler which directs the received signal into a waveguide terminated by a hard short and has a controllable shorting plane spaced approximately one-quarter wavelength from the termination point. The shorting planes are controlled by the output data signals and each introduces a 180 degree phase shift in the respective signal component when activated. The reflective couplers direct the selectively phase shifted signal components to an in-phase combiner, where they are combined to produce the quadrature phase shift keyed output signal.

TECHNICAL FIELD

This invention is related to a high frequency transmitter. Moreparticularly, this invention relates to a high frequency quadraturephase shift keying transmitter implemented in a waveguide.

BACKGROUND OF THE INVENTION

Evolving high frequency communications technologies operating fromaround 27 GHz to 31 GHz are generally full-duplex in nature and aredeployed in terrestrial microcells or configured to directly exchangedata with low earth orbit satellites. Service providers typically dividethe allocated transmit and receive frequency spectrums so that thetransmitter in the satellite or base station operates in a lower, andtherefore easier to work with, frequency block of around 27 GHz whilethe transmitter in the consumer's device operates at the higherfrequency, e.g. 31 GHz.

FIG. 1 is a high level diagram of a conventional quadrature phase shiftkeying (QPSK) transmitter 10 used to transmit at around 31 GHz. Thetransmitter consists of a channel oscillator 11 operating at anintermediate frequency (IF), e.g., 1 GHz, which feeds a quadraturedivider 12 that produces two output signals having a relative phasedifference of 90 degrees. These signals are provided to separatebi-phase modulator switches 13, 14 which output either the originalinput signal, or the input signal phase shifted by 180 degrees,depending on the value of a respective input data bit 15, 16. Thus, thefirst modulator 13 will produce an output signal with a phase of 0degrees or 180 degrees and the second modulator 14 will produce a signalwith a phase of 90 degrees or 270 degrees, according to the value of theinput data bits 15, 16. The output signals of the modulator switches aremixed with an in-phase combiner 17 to produce a four vector outputsignal 18.

The intermediate frequency QPSK signal 18 is input to a preamplifier 19via a coaxial cable. The amplified signal is processed by an imagefilter 20 to reduce noise and then combined with a signal from a localhigh frequency oscillator 21 operating at the transmit frequency, e.g.,31 GHz, with a mixer 22. A local trap 23 may also be utilized to clipout emissions outside a particular bandwidth in order to comply withapplicable government regulations. The final signal is then input to thetransmit amplifier 24.

It is very difficult to economically generate enough power at 31 GHz touplink a signal to a satellite or base station. A conventional solidstate 31 GHz transmit amplifier is produced as a thin film integratedcircuit using GaAs technology. Devices of this type which are powerfulenough to produce a one-watt output signal typically cost severalthousand dollars each. The high cost of the amplifier places the totalcost of the transceiver unit, which includes a transmitter, a receiver,an antenna, and the equipment housing, well beyond the price range ofmost interested consumers.

Accordingly, it is an object of the invention to provide a high powertransmit block for consumer satellite uplinks which may be inexpensivelymanufactured.

It is a further object of the invention to provide a high frequencytransmitter which eliminates the need to internally generate a modulatedsignal at an intermediate frequency before producing the high frequencyoutput signal.

Yet another object of the invention is to provide a high frequency QPSKtransmitter block in which the signal modulation is implemented in awaveguide.

SUMMARY OF THE INVENTION

According to the invention, two Gunn diode cavity oscillators areemployed as a quadrature signal source. The cavity oscillators areconfigured to operate at the transmit frequency, e.g., 31 GHz. The firstoscillator is driven by steering voltage which is generated by a phaselocked loop. A feedback path from the first oscillator to the PLL isprovided to maintain the oscillation phase. The second cavity oscillatoris slaved to the first at a specified phase vector, preferably bymagnetically linking the oscillators through integral wall aperturesdesigned to ensure a frequency coherence between the two oscillatorswhile maintaining the specified phase vector over the entire normalfrequency bandwidth of the devices. Each cavity oscillator is coupled toa waveguide to produce an output signal. The coupling point is selectedso that the output signals, i.e., quadrature vectors, are 90 degrees outof phase with each other.

The quadrature vectors are presented to a mirrored pair of bi-phase,solid-state switches realized in a waveguide. Each switch is preferablycomprised of a magnetic reflective coupling structure which can beswitched according to the value of an input data bit between a hard-wallwave guide short and a compensated, electrically generated shortingplane, such as a diode placed within the waveguide. The distance betweenthe switchable shorting plane and the hard-wall short is selected toproduce a switchable net phase change of 180 degrees. The waveguideoutput from each bi-phase switch is connected to an in-phase combiner toyield a combined QPSK signal that can be applied directly to an antenna.

Accordingly, the present invention allows the generation and combinationof two quadrature signal sources, each independently bi-phased switched,in a waveguide system which accepts input data directly and produces aQPSK output signal at the transmit frequency without needing anintermediate frequency stage. Further, the transmitter of the presentinvention eliminates the majority of components of a conventionaltransmitter. In particular, the power for the output signal is supplieddirectly from the cavity oscillators and coupled to the transmitteroutput through efficient waveguide structures. Therefore, separate highfrequency signal amplifiers are not required.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other features of the present invention will be morereadily apparent from the following detailed description and drawings ofillustrative embodiments of the invention in which:

FIG. 1 is a block diagram of a conventional high frequency QPSKtransmitter;

FIG. 2 is a block diagram of a high frequency QPSK transmitter accordingto the present invention;

FIG. 3a is top cross-sectional view of one embodiment of the transmitterof FIG. 2;

FIG. 3b is a cross-sectional view of the cavity oscillators along line3B--3B; and

FIG. 4 is a top cross-sectional view of a second embodiment of thetransmitter of FIG. 2.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

Turning to FIG. 2, there shown is a block diagram of a high frequencyQPSK transmitter 30 according to the present invention. Two Gunn diodecavity oscillators 32, 34 are each configured to operate at the transmitfrequency, e.g., 31 GHz. The first oscillator 32 is phase locked to theselected channel frequency and produces an output signal in waveguide40.

Preferably, locking is accomplished by applying a steering voltage 36 tothe oscillator 32. The steering voltage 36 is generated by a phaselocked loop ("PLL") 38 which preferably operates at only a fraction ofthe selected cavity oscillation frequency. A feedback path 43 taps thesignal in waveguide 40 and provides it to the PLL, 38 to maintain theoscillation phase.

The second cavity oscillator 34 is connected to the first oscillator 32in a manner which slaves the oscillation of the second oscillator 34 tothe first oscillator 32. In the preferred embodiment, the secondoscillator 32 is magnetically linked to the first oscillator 32 at aspecified phase vector through integral wall slots and a couplingaperture 45. This arrangement permits synchronizing energy from thefirst oscillator 32 to travel into the second oscillator 34 where itfunctions as a steering signal which synchronizes the phase andfrequency of oscillation of second oscillator 34 to that of the firstoscillator 32. Thus, the two oscillators can be controlled from a singleadjustment point. Preferably, the slots and aperture 45 are designed toensure a frequency coherence between the two cavity oscillators 32, 34,while maintaining a specified phase vector between the oscillators 32,34 over the entire normal frequency bandwidth of the devices.

Other slaving arrangements known to those of skill in the art may alsobe used. For example, the steering voltage 36 from PLL 38 may be used todrive the second oscillator 34. Alternatively, another PLL may beutilized to drive the second oscillator 34, which PLL is synchronized tothe first PLL 38.

Each cavity oscillator 32, 34 is coupled to a respective outputwaveguide 40, 42 to supply an output vector 44, 46. The oscillators 32,34 preferably have a cavity configuration which provides for outputvoltage signals 44, 46 extracted at different points to have differentphases. By selecting different extraction points for two oscillators 32,34, the output signals 44, 46 will be out of phase. The coupling pointsbetween the output waveguides 40, 42 and the oscillators 32, 34 areselected so that the two output vectors 44, 46 are 90 degrees out ofphase with each other. The extraction points may be adjusted as requiredto compensate for any phase differences introduced by the couplingmethod. According to the invention, these output signals are employed asa quadrature signal source.

Each quadrature vector 44, 46 is presented to a bi-phase, solid-stateswitch 48, 50. The configuration of the waveguides 40, 42 between theoscillators 32, 34 and the switches 48, 50 is chosen so that thewaveguides 40, 42 have substantially the same electromagnetictransmission characteristics so that any phase shifts which areintroduced are introduced equivalently to the generated signals 44 and46. This preserves the phase relationship between the two signals in amanner which is independent of the frequency of oscillation. Preferably,the waveguides 40, 42 are substantially symmetrically identical, i.e.,the have substantially the same shape, or are rotations and/or mirroredversions of each other, so that generated signals 44, 46 travel the samedistance along the same shape of path.

According to the invention, each bi-phase switch 48, 50 is realized in awaveguide and is comprised of a magnetic reflective coupling structureconnected to a waveguide which can be switched according to the value ofan input data bit between a hard-wall wave guide short and one or morecompensated, electrically generated shorting planes. The distancebetween the switchable shorting planes and the hard-wall short isselected to produce a switchable net phase change of 180 degrees, takinginto consideration any phase shift introduced by the parasiticcapacitance of the shorting switch in the off state. Thus, the output 52of the first bi-phase switch 48 will have a phase of either zero or 180degrees and the output 54 of the second bi-phase switch 50 will have aphase of either 90 degrees or 270 degrees, depending on the states ofthe switches 48, 50 as selected by the input data.

The output vectors from switches 48, 50 are passed through waveguides52, 54 which are connected to a conventional in-phase combiner 56. Thecombiner 56 produces a combined QPSK signal 58 which can be applieddirectly to a broadcast antenna 60.

According to the invention, virtually the entire signal path between theoscillators and the antenna is a waveguide structure. There are nointermediate stages in which the signal is converted from one frequencyto another. Instead, the signals originally generated by the oscillators32, 34 are the ones which are ultimately output and transmitted. Asignificant advantage of this arrangement is that the output power ofthe transmitter 30 is supplied directly from the oscillators 32, 34 andlimited only by the efficiency with which the signals are passed by thewaveguide structures.

Turning to FIG. 3a, there is shown a top cross-sectional view of oneembodiment of the transmitter 30 of FIG. 2. According to the invention,the entire transmitter apparatus is provided as a waveguide structurehaving three primary elements: a quadrature vector source 110, a phaseswitching assembly 112, and an in-phase combiner 114, shown hereseparated by lines 84 and 100. Preferably, conventional waveguide, suchas WR-28, and coupling aperture arrangements are utilized throughout.

The cavity oscillators 32, 34 are coupled to respective waveguides 40,42 through coupling apertures 62, 64. The apertures 62, 64 arepositioned on the oscillators 32, 34 so that the signals entering eachof the waveguides 40, 42 are substantially 90 degrees out of phase witheach other.

Preferably, the two cavity oscillators 32, 34 have a 0-1-0 cavityconfiguration which advantageously allows output signal vectors to beextracted at different points along the cavity to thereby providedifferent output signal phases. Thus, two cavities oscillatingsynchronously with each other can produce a pair of output vectors withany desired relative phase relationship. Because the phase of the outputsignal depends on the physical location of the extraction point, thephase relationship between the two signals remains substantiallyconstant with changes in the oscillation frequency.

As shown in FIG. 3a, the circumferential position of the couplingaperture 62 between the first oscillator 32 and the waveguide 40 issubstantially 90 degrees from the position of the coupling aperture 64between the second oscillator 34 and the waveguide 42. In the preferred0-1-0 cavity configuration, this arrangement provides the desired 90degrees phase difference.

A cross section of the cavity oscillators 32, 34 along line 3B--3B isshown in FIG. 3b. Eachzavity oscillator 32, 34 contains a respectiveGunn diode 66, 68 coaxially aligned with the axis of the respectiveoscillator cavities. DC power for the diodes 66, 68 is supplied througha coaxial cable 70, 72. The coaxial cable 70 is also used to provide thesteering voltage signal 36 from the PLL 38 to the first oscillator 32.Each oscillator 32, 34 has a respective coupling slot 74, 76 which isconnected to a coupling aperture 78 to connect the two oscillators 32,34 as described above. Also shown in FIG. 3b is a coupling slot 80 inthe second oscillator 34 which connects it to the waveguide 42.

Returning to FIG. 3a, each waveguide 40, 42 connects a respectiveoscillator 32, 34 to one of the bi-phase switches 48, 50. The length andconfiguration of each connecting waveguide 40, 42 is selected so thatthe phase difference between the two signals is preserved. In thepreferred embodiment, the connecting waveguides 40, 42 are substantiallymirror images of each other. This ensures that both waveguides 40, 42will introduce the same phase shift, thus preserving the phaserelationship, and will also have the same degree of attenuation, therebykeeping the signal strength balanced.

The signals received from the waveguides 40, 42 are preferably directedinto the switching portions of the phase switches 48, 50 by reflectivecoupling structures 49, 51. These coupling structures 49, 51 are alsoconfigured to properly direct the phase-switched outputs 52, 54 from theswitches 48, 50 into the in-phase combiner 56.

Each switch 48, 50 is preferably realized in a waveguide whichterminates in a hard reflecting short 86, 88 and has a shorting diode90, 92 placed in the signal path (shown extending into the plane). Thediodes 90, 92 function as electrically variable shorts which act asswitching points, effectively altering the length of the respectiveswitch waveguide 48, 50 when they are conducting.

Ideally, a phase shift of 180 degrees is provided when the diodes areconducting and are placed one-quarter wavelength from the hard shortingplane. However, placing a diode in the waveguide introduces a parasiticcapacitance which may alter the phase of the signal as it passes throughthe off-state diode while traveling to and from the hard short 86, 88.(When the diode is conducting, it functions as a short and the parasiticcapacitance is of little concern). Because only a relative phase shiftis required, the position of the diode is adjusted to compensate for theintroduced phase shift.

Those skilled in the art will recognize that any type of mirroredwaveguide switching arrangement may be utilized with the signal sourcediscussed above and that various different waveguide structures may beused to provide the preferred hard and diode shorting points. In theembodiment shown in FIG. 3a, the impedance of the waveguide is loweredby adding ridges 96, 98. Preferably, a double-ridged waveguideconfiguration is used. This configuration focuses the magnetic field ofthe applied signal to directly impact the solid-state switch point whichforms the shorting plane, increasing the efficiency of the switch.

As with the connecting waveguides 40, 42, the two switches 48, 50 arepreferably substantially mirror images of each other. This ensures thatany inherent phase shifts which are introduced by the switchingstructure are equally represented in both signals and therefore cancelout.

The (phase-shifted) output of the switches 40, 50 are applied to anin-phase combiner 56 to produce a QPSK output signal 58. Combining theoutput of the quadrature vectors in this manner advantageously providesa power-doubling effect in the combined signal with regards to signalamplitude without distortion.

An alternative, and more preferred arrangement of the transmitter 30 isshown in FIG. 4. The overall configuration is the same as shown in FIG.3a. However, instead of using a single-point reflective switchingstructure as the bi-phase switch, quadrature switching structures 140,142 are utilized. The switching waveguides 140, 142 are each comprisedof two balanced waveguides 144, 146, 148, 150, each having its ownswitch point 152, 154, 156, 158. A balanced quadrature structure is moreefficient than the single-point reflective structure of FIG. 3a. Inaddition, the use of two switch points isolates the switching andreduces interference.

An alternate configuration 160 for the in-phase combiner is also shown.This configuration has different reflection points than the combiner 56shown in FIG. 3a and a lower signal loss. Also provided are steps 162which may be used to match the impedance of the waveguide at the outputto that of the transmit antenna structure.

Various modifications may be made to the transmitter structure describedabove without departing from the scope of the invention. For example,more than two oscillators may be slaved together and used to produceoutput vectors having phase relationships other than 90 degrees. Fouroscillators may be provided and output signals selected to have phaserelationships of 0, 45 degrees, 90 degrees, and 135 degreesrespectively. Each output signal could then be supplied to a bi-phaseswitch as described above and the results merged with a 4-input bi-phasecombiner to thereby allow four data bits to be simultaneouslytransmitted as an eight data-point constellation.

Additional pairs of switches which provide a phase shift other than 180degrees may also be introduced to increase the data carrying capacity ofthe structure. For example, by placing the shorting diode(s) atapproximately 1/8 wavelength from the hard short, a 45 degree phaseshift may be selectively introduced. Adding a mirrored pair of theseswitches to the QPSK structure shown above allows an 8-point signalconstellation to be produced. Alternatively, multiple short points maybe introduced in a single switch to provide for several selectable phaseshifts.

The oscillators 32, 34 and connecting waveguides 40, 42 may further beutilized as a signal source independent of the transmitter arrangement30 described above. Thus, for example, a signal source 110 havingoutputs 80, 82 (along dividing line 84) can be provided without theremaining switching structure. By varying the point at which theoscillators 32, 34 are tapped and/or varying the configuration of thewaveguides 40, 42, a dual-vector source can be produced with any desiredphase relationship. Advantageously, the phase relationship remainssubstantially constant even as the frequency of oscillation is changed.Various applications for such a stable signal source will be apparent tothose skilled in the art. For example, the power level of theoscillators may be modulated, the output polarized, and a combinerutilized to create a circularly polarized, amplitude modulated outputsignal for use in satellite and radar applications or the like.

While the invention has been particularly shown and described withreference to preferred embodiments thereof, it will be understood bythose skilled in the art that various changes in form and details may bemade therein without departing from the spirit and scope of theinvention.

I claim:
 1. A phase-shift keying transmitter comprising:a signal sourceproviding first and second signal components having substantially thesame frequency and being out of phase with each other; a first waveguidereceiving said first signal component; a second waveguide receiving saidsecond signal component; a first phase modulator connected to said firstwaveguide and responsive to a first control signal to selectivelymodulate the phase of said first signal component by a first predefinedamount; a second phase modulator connected to said second waveguide andresponsive to a second control signal to selectively modulate the phaseof said second signal component by a second predefined amount; and anin-phase combiner connected to said first and second phase modulators soas to combine said selectively modulated first and second signalcomponents and provide a phase-shift modulated combined output signal.2. The transmitter of claim 1, wherein said first and second signalcomponents are substantially 90 degrees out of phase.
 3. The transmitterof claim 2, wherein said first and second predefined amounts aresubstantially 180 degrees.
 4. The transmitter of claim 1, wherein saidsignal source comprises a first oscillator providing said first signalcomponent and a second oscillator providing said second signalcomponent.
 5. The transmitter of claim 4, wherein said first and secondoscillators comprise first and second cavity oscillators configured tomaintain a constant relative phase relationship.
 6. The transmitter ofclaim 5, wherein:said first and second oscillators are magneticallycoupled to each other.
 7. The transmitter of claim 5, each of said firstand second oscillator comprising a cavity wall having a wall aperturetherein;a coupling aperture connected between the wall aperture in saidfirst and second oscillators to phase couple said oscillators to eachother.
 8. The transmitter of claim 4, further comprising a phase lockedloop driving at least one of said first and second oscillator.
 9. Thetransmitter of claim 1, wherein said first and second waveguides havesubstantially identical electromagnetic transmission characteristics.10. The transmitter of claim 9, wherein said first and second waveguidesare substantially symmetrically identical.
 11. The transmitter of claim1, wherein each said phase modulator comprises:a modulating waveguideterminated by a reflecting short; and a selectively engagable shortingplane within said modulating waveguide responsive to said respectivefirst or second control signal, the distance between said shorting planeand said reflecting short determining the degree of selective phasemodulation.
 12. The transmitter of claim 11, wherein said shorting planecomprises a diode placed within the modulating waveguide, saidrespective control signal selectively biasing said diode on or off. 13.The transmitter of claim 11, wherein each said modulating waveguidecomprises a double-ridged waveguide.
 14. The transmitter of claim 11,wherein each said phase modulator further comprising a magneticallyreflective coupler connected between said respective first or secondwaveguide, said modulating waveguide, and said combiner, said couplerdirecting said respective first or second signal component from saidrespective first or second waveguide into to said modulating waveguideand directing said respective selectively modulated first or secondsignal component from said modulating waveguide into said combiner. 15.The transmitter of claim 11, wherein the distance between each saidshorting plane and respective reflecting short is approximatelyone-quarter of the wavelength of said signal components.
 16. Thetransmitter of claim 1, wherein each said phase modulator comprises apair of balanced waveguides, each one of said pair of balancedwaveguides terminating in a reflecting short and having a selectivelyengagable shorting plane responsive to said respective first or secondcontrol signal, the distance between said shorting planes and saidrespective reflecting shorts determining the degree of selective phasemodulation.
 17. The transmitter of claim 1, wherein said in-phasecombiner comprises a waveguide.
 18. A method of generating a phase-shiftkeyed signal comprising the steps of:providing first and second signalcomponents having substantially the same frequency and being out ofphase; coupling said first and second signal components to first andsecond phase modulators through first and second waveguides; in responseto first and second control signals, selectively modulating the phase ofsaid first and second signal components; and combining the modulatedoutput of said first and second modulators with an inphase combiner tothereby provide said phase-shift keyed signal.
 19. A phase shift keyingtransmitter comprising:a signal source comprising first and secondcavity oscillators producing respective first and second signalcomponents having substantially the same frequency and beingapproximately 90 degrees out of phase; a phase switching assembly havingfirst and second substantially symmetric biphase switches receiving saidfirst and second signal components respectively, and providing first andsecond switched output signals, said first and second switchesselectively switching the phase of said respective output signal bysubstantially 180 degrees in response to a respective first or secondcontrol signal; and an in-phase combiner receiving said first and secondswitched output signals and providing a combined quadrature phase shiftoutput signal.
 20. The transmitter of claim 19, whereinsaid first andsecond cavity oscillators are Gunn diode cavity oscillators magneticallycoupled to each other; said cavity oscillators each have a 0-1-0 cavityconfiguration; said first signal component is extracted from said firstoscillator at a first coupling aperture; said second signal component isextracted from said second oscillator at a second coupling aperture;said second coupling aperture is substantially perpendicular to saidfirst coupling aperture.
 21. The transmitter of claim 20, furthercomprising:a first waveguide connected between said first couplingaperture and said first switch; a second waveguide connected betweensaid second coupling aperture and said second switch; said first andsecond waveguides being substantially electrically equivalent.
 22. Thetransmitter of claim 20, wherein each said bi-phase switch comprises:adouble ridged waveguide terminating in a hard short; and a diodepositioned within said waveguide approximately one quarter wavelength ofsaid first and second signal component, respectively, from said hardshort and being controlled by a control signal.
 23. The transmitter ofclaim 20, wherein each said bi-phase switch comprises:a first switchingwaveguide terminating in a first hard short; a first a diode positionedwithin said first waveguide approximately one quarter wavelength of saidfirst signal component from said first hard short and being controlledby a control signal; a second switching waveguide terminating in asecond hard short and balanced with respect to said first switchingwaveguide; a second a diode positioned within said second waveguideapproximately one quarter wavelength of said second signal componentfrom said second hard short and being controlled by said control signal.24. A phase shift keying transmitter comprising:first and second cavityoscillators generating first and second signal components havingsubstantially the same frequency and being out of phase; means forselectively modulating the phase of said first and second signalcomponent in response to first and second control signals, respectively;means for coupling said means for generating to said means forselectively modulating; means for combining said selectively modulatedfirst and second signal components to produce a phase-shift keyedsignal.
 25. The transmitter of claim 24, further comprising means toslave said second oscillator to said first.
 26. A phase shift keyingtransmitter comprising:a signal source producing first and second signalcomponents having substantially the same frequency and beingapproximately 90 degrees out of phase; a phase switching assembly havingfirst and second substantially symmetric biphase switches receiving saidfirst and second signal components respectively, and providing first andsecond switched output signals, said first and second switchesselectively switching the phase of said respective output signal bysubstantially 180 degrees in response to a respective first or secondcontrol signal, each said bi-phase switch comprising a double ridgedwaveguide terminating in a hard short, and a diode positioned withinsaid waveguide approximately one quarter wavelength of said respectivesignal component from said hard short and being controlled by therespective control signal; and an in-phase combiner receiving said firstand second switched output signals and providing a combined quadraturephase shift output signal.
 27. The transmitter of claim 26, wherein saidsignal source comprises magnetically coupled first and second Gunn diodecavity oscillators.
 28. The transmitter of claim 27, wherein:said cavityoscillators each have a 0-1-0 cavity configuration; said first signalcomponent is extracted from said first oscillator at a first couplingaperture; said second signal component is extracted from said secondoscillator at a second coupling aperture; and said second couplingaperture is substantially perpendicular to said first coupling aperture.29. The transmitter of claim 28, further comprising:a first waveguideconnected between said first coupling aperture and said first switch; asecond waveguide connected between said second coupling aperture andsaid second switch; said first and second waveguides being substantiallyelectrically equivalent.